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A4401 Schematic ( PDF Datasheet ) - Allegro Micro Systems

Teilenummer A4401
Beschreibung Automotive Low Noise Vacuum Fluorescent Display Power Supply
Hersteller Allegro Micro Systems
Logo Allegro Micro Systems Logo 




Gesamt 17 Seiten
A4401 Datasheet, Funktion
www.DataSheet4U.com
A4401
Automotive Low Noise Vacuum
Fluorescent Display Power Supply
Features and Benefits
Multiple output regulator
7 to 40 V input supply
Low EMI conducted and radiated emissions
Adaptive quasi-resonant turn on/off control
Minimal number of external components
Enable input which can be driven with respect to the
battery voltage
Package: 8-pin narrow SOIC (suffix L)
Description
This device provides all the necessary control functions to
provide the power rails for driving a vacuum fluorescent
display (VFD) using minimal external components. The power
supply is based on a quasi-resonant, discontinuous flyback
converter, operating near the critical conduction boundary. A
novel adaptive turn-on control scheme is used to optimize the
turn-on and turn-off phase of the MOSFET, to reduce EMI
emissions while minimizing switching losses.
The converter is self-oscillating, operating at switching
frequencies depending on the input voltage, load, and external
components. An onboard linear regulator that is powered
directly from the battery provides the housekeeping supply,
avoiding the need for complex bias supplies.
Internal diagnostics provide comprehensive protection
against overloads, input undervoltage, and overtemperature
conditions.
The A4401 is supplied in an 8-pin narrow SOIC package
(suffix L), which is lead (Pb) free, with 100% matte-tin
leadframe plating.
Approximate Scale 1:1
Typical Application
+VBAT
ECU
VIN
A4401
LX
GD
ISS
EN
GND
VA
COMP
0V
0V
VFD
A4401-DS






A4401 Datasheet, Funktion
www.DataSheet4U.com
A4401
Automotive Low Noise Vacuum
Fluorescent Display Power Supply
In the event of an overload, the current demand signal
produced by the Gm amplifier restricts the output cur-
rent by introducing pulse-by-pulse current limiting.
Regulation Voltage
The feedback resistors, R5 and R6, determine the
voltage of the output rail to which they are connected,
according to the following formula:
VOUT =
VREF ×(R5 + R6)
R6
,
(1)
where R6 should be approximately 5 kΩ.
The internal 1.2 V reference has a ±2% worst case
tolerance, plus there is an input bias current, IBIAS, on
the feedback node, VA, that has a small influence. This
current flows into the ground referenced resistor, R6,
creating a small voltage offset.
In applications where the main control output (anode
or grid) can run at relatively light loads (relative to the
filament load), it may be necessary to “mix” the feed-
back signal. This involves adding an additional feed-
back from the filament output to the VA input. Note
that this only applies to DC filament outputs.
R3 Current Sense Resistor Selection
To determine the resistance value, the maximum peak
current needs to be determined. First determine the
average input current, IAV , as follows:
IAV =
POUT(max)
H% × VIN(max)
,
(2)
where POUT is the output power. Then, the peak cur-
rent through the sense resistor:
IPK
=
2 × IAV
D(max)
,
(3)
where D is limited to 0.7, or can be precisely found as
described in the Magnetics Design section. Note that
a D of 0.7 is chosen in order to achieve 0 V switching
with a VBAT of 13.5 V.
To determine the sense resistor value, assume that the
minimum sense voltage before current limiting occurs
is 600 mV. A reasonable maximum voltage to select
during normal operation would be 500 mV. Then, the
resistor value can be found as follows:
RSENSE
=
500 mV
IPK
.
(4)
The power losses in the resistor can be found by first
determining the rms current through it:
IRMS =
IPK ×
⎜⎜
D(max)
3
½
⎟⎟
.
(5)
Then, the losses in sense resistor are:
PRDS = I ²RMS × RSENSE .
(6)
The power rating of the resistor can be selected based
on the power dissipation. When selecting a resistor
it is worth noting that the maximum power rating
is valid up to 70°C and derates linearly to 0 W at a
temperature of typically between 120°C and 140°C.
Check the resistor manufacturer guidelines.
Note that is imperative that this resistor be a low
inductance type; avoid wire wound. Standard surface
mount devices are usually acceptable.
Soft Start
When power is initially applied, assuming the input
voltage turn-on threshold is reached, and the EN input
is enabled, the controller is initiated and the MOSFET,
Q1, is turned on for the first switching cycle. Initially,
while the output volts are rising towards the target reg-
ulation point set by the soft start circuit, the MOSFET
will run at current limit.
During a soft start cycle, the reference voltage is
ramped from 0 to 1.2 V in 32 steps over a period of 10
ms under the control of a DAC. This forces the output
of the amplifier to vary between 0.8 and 1.5 V, which
in turn reduces the effects of inrush current and volt-
age overshoot on the outputs.
Allegro MicroSystems, Inc.
6
115 Northeast Cutoff, Box 15036
Worcester, Massachusetts 01615-0036 (508) 853-5000
www.allegromicro.com

6 Page









A4401 pdf, datenblatt
www.DataSheet4U.com
A4401
Automotive Low Noise Vacuum
Fluorescent Display Power Supply
where IOUT is the maximum load current, and D'(max)
is the duty cycle, limited to 0.3.
Then the rms current in the winding is:
IRMS
=
IPK ×
⎜⎜
D'
3
½
⎟⎟
,
(32)
Because the number of turns has already been worked
out, the ampere-turns factor can now be determined.
After all of the ampere-turns are known for each wind-
ing, the bobbin window can be apportioned to each
winding. It is recommended that the current density in
each winding should be kept below 5 A per mm2.
Another consideration when selecting the wire gauge
is the skin depth (depth within which the current
flows), especially at higher frequencies. Skin depth
can be calculated as:
δ=
75
f
½
SW
,
(33)
For example, if 45 kHz were the minimum frequency
at minimum input voltage and maximum load, then
to ensure maximum wire utilization for the first four
switching harmonics, the switching frequency would
be 180 kHz. The conduction depth at 180 kHz would
equal 0.18 mm, therefore, the wire diameter should
not exceed 0.36 mm. For any particular winding, if the
current rating of the wire is insufficient even though
the wire meets the skin depth criteria, multiple wires
wound in parallel will be necessary.
It is recommended to locate the start and finish of
each winding as close as possible on the bobbin. This
minimizes the “loop area” and reduces the effects of
noise pick-up.
C11 Resonant Capacitor Selection
The resonance that occurs when the MOSFET, Q1,
turns off is formed by the interaction of the primary
magnetizing inductance and the capacitance between
the LX node (the drain terminal of the MOSFET) and
ground. The design is optimized for a half resonant
period of 1 μs. This means the resonant capacitor
value can be found from the following formula:
CRES = ⎜⎜⎛TR
⎟⎟
²
×
1
LPRI
,
where TR is a half resonant period of 1 μs.
(34)
It is advisable to measure the half resonant period in
the application, as the parasitic capacitance between
the LX node and ground can be substantial and may
even be sufficient to meet the requirements with very
little additional capacitance.
PCB Layout Guidelines
The layout can be considered as two blocks: primary
and secondary:
Primary Block To minimize parasitic noise appearing
on the ground return, and at the LX and ISS nodes,
as well as to maximize the effectiveness of the EMI
VBAT
CIN
LPRI
A4401
LX node
MOSFET
Q1
ISS node
RSENSE
When winding the high voltage windings, such as for
the anode or grid, it is advisable to insert a layer of
polyester insulating tape between each layer as well as
between adjacent windings.
Figure 3. Main power loop
Minimize this loop area
Allegro MicroSystems, Inc.
12
115 Northeast Cutoff, Box 15036
Worcester, Massachusetts 01615-0036 (508) 853-5000
www.allegromicro.com

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